Active RIAA Equalization

Let’s now look at an active RIAA equalization network. Below is the classic active feedback equalization topology.

 

R₂ X C₂ = 3180uS

R₁ X C₁ = 75uS

(R₁ || R₂) X (C₁ + C₂) = 318uS

GAIN_DC = (R + R₁ + R₂)/R

Gain_AC in dB= 20Log(GAIN_DC) - 20dB

The formulas look simple enough, until you actually take a hand calculator to them, as meeting the third line’s condition (while using readily available part values) is tough. Below is a simple RIAA phono preamp based on Analog Device’s excellent AD712 op-amps, which are known for their low noise and smooth sound. ( Active RIAA Op-amp Preamp.ckt  Ver 4.2) ( Active RIAA Op-amp Preamp.ckt  Ver 2000)

Below we see the frequency response of the circuit above.

The circuit improves upon the classic topology in that the second (the RIAA equalizing) op-amp runs in the inverting mode, which results in two benefits. The first is that the single coupling capacitor is enough to keep both the first amplifier’s DC offset from coupling to the second amplifier and to keep the second amplifier from amplifying its own input DC offset, as resistor R₅ does not terminate into a path that leads to ground, so it cannot define a voltage divider with resistors R₁ and R₂. (In other words, the second amplifier function as a unity-gain amplifier at DC, which prevents it from amplifying its own input DC offset.)

 The second benefit deriving from inverting amplifier operation is that the inverse RIAA equalization curve is more closely followed, as the equalization network is effectively grounded at the op-amp’s negative input; whereas in a non-inverting configuration, the equalization network will be thrown off by resistor R₅’s resistance. (Of course, the lower R₅’s resistance, the less the equalization’s deviation.) An added benefit to running this amplifier in the inverting mode is that the 75µS (2122Hz) low-pass filter portion of the RIAA equalization is accurately followed beyond the audio band, instead of flattening out due to the non-inverting amplifier’s gain hitting the floor of unity gain, creating the famous fourth pole.  

In other words, at extremely high frequencies, capacitors C₁ and C₂ become effectively dead shorts to the signal and resistor R becomes just a load resistor to be driven and the amplifier sees 100% of its output returned to its negative input, reducing its gain to unity, 1V in, 1V out. A simple RC low-pass filter is often added to an actively equalized non-inverting preamp to compensate for the RIAA curve hitting the unity gain floor. How do we determine the value for the added resistor and capacitor? We start at the DC gain and deduce the frequency at which AC gain drops to unity: 

            T₁T₂ / GAIN_DC = T₃T₄

            T₄ = T₁T₂ / (T₃GAIN_DC)

        \T₄ = 750µS/GAIN_DC

where T₁ = 3180µS, T₂ = 75µS, T₃ = 318µS, and GAIN_DC = DC gain. Translating this time constant into a –3dB frequency is easy enough: just divide 159154.9 by T₄ (as µS, not seconds). For example a DC gain of 1,000 (+40dB @ 1kHz) will put the flattening at roughly 212kHz. On the other hand, some argue that record itself (and/or the cartridge) adds the missing time constant in that after a few playings, it is unlikely that the frequency response extends beyond 25kHz. Below is an example circuit that holds a compensating low-pass filter.

 

Is phase inverted signal a problem? If you think it is, then it is. Fortunately, the solution is simple: just reverse the leads that connect to the phono cartridge, which will send an inverted signal to the preamp’s input, which in turn will be inverted again, resulting zero phase inversion (assuming the recording studio’s microphones and preamps and equalizers, mixing boards, maybe an analog-to-digital converter or two, and the cutting-head amplifiers all observed a consistent phasing).